Sensitivity and selectivity issues in crystal radio sets including diode problems; measurements of the Q of variable and fixed capacitors, RF loss in slide switches and loss tangent of various dielectrics
By Ben H.
Tongue
Part A: The Issues.
The sensitivity and selectivity in a crystal radio set can
be impacted by many factors, including:
- A1: Ineffective coupling of the antenna to the tank
circuit.
- A2: Resistive RF losses in the tank circuit.
- A3: Inappropriate diode.
- A4: Unintentional diode damage from exposure to electrostatic discharge (ESD) or voltages higher than the minimum reverse voltage rating. This is one cause of the loss of weak-signal sensitivity over time.
- A5: Relations between antenna-ground system,
diode RF input and tank loss resistance, as effecting selectivity
and loaded Q.
- A6: Audio transformer loss.
A1. There is a practical
minimum limit to the possible impedance transformation ratio of
the series resistance of the antenna-ground circuit to the shunt
value desired across the tank circuit, when the transformation
means is just a series capacitor between the antenna and the top
of the tank. This problem occurs when the capacitor required
for the desired impedance transformation becomes so large that
it causes the tank inductance to resonate below the desired frequency
(See Article #22, Part 2). The solution here is to use a
lower value inductance for the tank or to tap the antenna down
on the tank (towards ground). This is why some experimenters
find that low inductance tank circuits seem to work better than
those of higher inductance. If one does not use a series
capacitor for impedance transformation, the antenna may be just
tapped down on the tank. Another alternative is to connect
the antenna-ground system to a low value unturned inductance that
is coupled to the tank.
The main advantage of using a series capacitor
connected to the top of the tank (for impedance matching) is that
it moves the undesired short wave resonance of the antenna-ground
circuit (present in every single tuned crystal radio set) to the highest
possible frequency and reduces its strength. A disadvantage
is that unless a high enough Q variable capacitor is used, insertion
power loss is increased, especially at the high frequency end
of the band. See Part B, Section 1 of this Article and Article
#22, Part 7.
A2. Resistive RF losses in the
tank circuit are affected by: 1) Losses in any capacitor
used for tuning or RF coupling. 2) Physical size of the
coil and such items as length/diameter ratio, cross-section size
and shape, and turns spacing (to reduce coil proximity losses).
3) Loss tangent in the coil form material, wire insulation
and all dielectric material penetrated by the electric field of
the coil. 4) Wire size and plating, if any. Silver
plating is good but tin plating is bad, especially at the
high end of the band. 5) Wire construction such as litz,
solid or un-insulated stranded. The latter should be avoided.
6) Switches (if used). 7) Magnetic coupling from the coil
to nearby lossy metallic objects. 9) Capacity coupling from
"hot" high impedance RF points through a lossy RF return path
to ground. See Article #22, point 7 of Part 10. Some
comments: The loss from the loss tangent of the dielectric material
used for the mounting base of detector stands can be nontrivial.
Loss tangent of the material used for a front panel can cause dissipative
loss if terminals provided for the connection of an external diode
are too close together. See Part B, section 3 of this Article.
A3. A diode with too low an
axis-crossing resistance (too high a saturation current) will
resistively load the tank too heavily, causing a low loaded Q
that results in loss of selectivity and sensitivity.
A diode with too high an axis-crossing resistance (too low a saturation
current) will increase selectivity because it only lightly resistively
loads the tank circuit. The disadvantage is that sensitivity
is reduced (many people liken this to the diode having too high
a "turn-on" voltage.) and considerable audio distortion is generated.
A little reverse or forward DC voltage bias will usually fix up
these performance problems. See Article #9.
A way to check whether weak signal performance would
be improved if one used a diode with a different saturation current
(Is), but without experimenting with DC bias, is as follows: 1)
Give the diode a one second or so spray with an aerosol "component
cooler" . The reduction in temperature will temporarily
substantially reduce the Is of the diode. If performance
improves during the subsequent warm-back-up period, but before
reaching room temperature, the diode has too high a room temperature
Is. 2) Heat the diode by holding a hot soldering iron next
to it for 5 seconds or so or give it a quick blow from a hot hair
dryer. If performance improves during the subsequent cool-back-down
period, but before reaching room temperature, the diode has too
low a room temperature Is. Is, for the usual Schottky diode,
changes by about two times for each 10° C. temperature change.
Germanium diodes probably act the same. Aerosol component
coolers are available from most Electronics Distributors, including
Radio Shack.
A4. Semiconductor diodes are subject to damage from exposure to electrostatic discharge (ESD) and voltages higher than their reverse breakdown values. Easy-to-check-for damage shows up as an increased reverse current when tested with a reverse DC voltage close to or less than the specified minimum breakdown voltage, as compared to an undamaged diode. Other types of damage can also occur. The effect on detector performance can be anywhere from a mild to a very great reduction of weak-signal sensitivity. This type of damage has the effect of placing a resistor across the diode, which reduces tank Q and adds a parasitic, unneeded audio load resistance. If one has an old
VOM having a d'Arsonval moving-coil meter movement (not a digital
type) such as the Triplett 360, Weston 980 or Simpson 260, one can do a quick check of the back leakage
of a Schottky diode by measuring its back resistance on the X1,000 range. If the needle does not move, the diode has probably not been damaged from ESD.
This test does not apply to so-called zero-bias Schottky detector diodes because of their usual low reverse breakdown voltages. Undamaged germanium diodes normally have greater reverse leakage
(lower back-resistance) than the usual Schottky
detector diodes. This does not effect performance in crystal set applications. The germaniums seem less subject to the electrostatic damage problem precisely because their lower back-resistance (and high reverse breakdown voltages) tends to bleed off any electrostatic charge that might accumulate, maybe from handling. Diodes with the lowest leakage
can be selected with a simple test as follows: Connect a 3-4.5 volt DC source, a 4.7k to 10k resistor, the diode and a
DVM set to read DC current in series. Polarize the battery
so that the diode is back biased. If the current is 2 uA
or less, the parasitic back leakage resistance is greater than
2 Megohms or so and all should be well, as far as weak signal loss is concerned.
The resistor is used to prevent damage to the test diode if it
is accidentally connected in the forward direction. This test may be used to sort out the best weak-signal sensitivity diodes and exclude damaged ones. It is probable that for a selection of diodes having the same part number, ones with the lowest reverse leakage current
will deliver the best weak-signal performance.
I believe poor weak-signal results some people have reported with the Avago (formerly Agilent) 5082-2835 and HSMS-2820 diodes can be laid at the doorstep of ESD damage. I experienced my first problem with these diodes by storing a few new ones in a propylene vitamin pill bottle for a while. When I wanted a new one I picked up the bottle, shook it (I think,) to see if something was inside and withdrew a diode. It performed badly, so I checked its back resistance with my Triplett 360 VOM as suggested above and found a low back resistance reading. Checking all the others, I found they were all bad. Static electricity did them in... beware! It was pretty stupid of me to not use anti-static bags. BTW, I've never had problems with those diodes if stored in paper bags. The bags must have had enough moisture content to provide sufficient conductivity to bleed off any static charge that might accumulate.
It is difficult to properly impedance match these diodes (see Part 4 of Article #5) because of their low saturation current (high axis-crossing resistance). They work best if 3 or or so are connected in parallel to raise the effective saturation current. If one wishes to use only one of these diodes, it could have a low forward bias applied to get the same result (see Article #9). Keep in mind that these diodes do not do very well on strong signals when optimally impedance matched for weak signal reception because of their low reverse voltage breakdown rating. Avago specifies that rating as 15 volts for these diodes. If a rectified voltage approaching 7.5 volts appears across the DC load (now operating in 'peak-detector' mode), a maximum reverse voltage of about 15 volts will appear across the diode for every cycle of the RF carrier. When the diode is operated at this signal level or greater, a 'bucking' rectified current flows through the diode reducing audio output and causing audio distortion. This problem is eliminated by using diodes having a greater reverse breakdown voltage such as most ITT FO 215 and most 1N34A units.
A5. A question asked by many
designers is this: What is the best approach when deciding how
to 'impedance match' the transformed antenna-ground-system resistance
to the RF input resistance of the diode detector? This would
be a no-brainer if the resonant resistance of the tank was infinite,
but it is not.
Table 1 - Definitions of Terms
|
IPL
|
Insertion power loss
|
Lo
|
Inductance of the tank
|
Ql
|
Loaded Q of the tank |
Qo
|
Unloaded Q of the tank |
Rag
|
Actual antenna-ground-system resistance |
Ragt
|
A antenna-ground-system resistance transformed
its value at the top of the tank |
Rd
|
Actual input RF resistance of the
diode. |
Rdt
|
Input RF resistance of the diode
as transformed to its value at the top of the tank. |
Ro
|
Resistor representing all losses in the tank |
Xa
|
Antenna-ground-system impedance transformation. |
Xd
|
Diode input RF resistance transformation. |
The presence of finite tank loss increases IPL from
the theoretical zero of the matched impedance case. It also reduces
selectivity. Fig. 1a shows a simplified schematic of a single-tuned
crystal radio set. The input impedance transformation Xa might
take the form of capacitor in series with the 'Rag' and the 'induced
voltage source' series combination or a tap on the tank. The
diode input RF resistance transformation might take the form of
a tap on the tank or a capacitor in series with the diode, along
with other components. See Son of Hobbydyne and Hobbydyne
II at http://www.hobbytech.com/crystalradio/crystalradio.htm.
Table 2 shows calculated data for the simplified schematic
shown in Fig.1b.
Table 2 - Calculated
Insertion power loss (IPL), ratio of loaded to unloaded Q and
impedance match (return loss*) in a single-tuned circuit.
|
Line #
|
Ragt/Ro
|
Rdt/Ro
|
IPL
in dB
|
Ql/Qo
|
Input match: S11 in dB
|
Output match:
S22 in dB
|
1
|
0.250
|
0.250
|
1.023
|
0.1111
|
-19.85
|
-19.85
|
2
|
0.333
|
0.500
|
1.761
|
0.1667
|
-infinity
|
-9.53
|
3
|
0.667
|
0.667
|
2.50
|
0.250
|
-12.04
|
-12.04
|
4
|
1.000
|
0.500
|
3.01
|
0.250
|
-6.02
|
-infinity
|
5
|
0.500
|
1.000
|
3.01
|
0.250
|
-infinity
|
-6.02
|
6
|
1.000
|
1.000
|
3.52
|
0.333
|
-9.54
|
-9.54
|
7
|
2.000
|
0.667
|
4.77
|
0.333
|
-3.52
|
-infinity
|
8
|
0.667
|
2.000
|
4.77
|
0.333
|
-infinity
|
-3.52
|
9
|
2.000
|
2.000
|
6.02
|
0.500
|
-6.02
|
-6.02
|
10
|
3.000
|
3.000
|
7.96
|
0.600
|
-4.34
|
-4.34
|
*Return loss is a measure of the "goodness" of an
impedance match. A value of minus infinity indicates a
'perfect impedance match' (all the available input power is
delivered to the load). A value of zero indicates a 'perfect
impedance mismatch' (all of the available input power is reflected
back to the source, and none is delivered to the load). An
intermediate value indicates the amount of available power that
is reflected back to the source by the load. The usual
way of referring to mismatch of a two-port network is by using
S parameters. S11 is the input reflection coefficient
and S22, the output reflection coefficient. The magnitude
of a voltage reflection coefficient is 20*log|{(Rload-Rsource)/(Rload+Rsource)}|.
One reference is Radiotron Designer's Handbook, Fourth
Edition, pp 891-892. In our case, consider out little circuit
(at resonance) to be a zero length transmission line having
attenuation. Note that the Handbook was written before 'S" parameters
were widely used.
Table 1 shows the tradeoff between IPL and selectivity. The
lower Rag and Rd become, the lower IPL becomes, but Ql/Qo drops
(poorer selectivity). Higher values of Rag and Rd result
in greater IPL and greater Ql/Qo (greater selectivity). Ramon
Vargas has suggested that many people consider the parameters
on line 4 to be close to a practical optimum, and they are.
Line 3 shows alternate parameters for achieving the same
selectivity at an IPL 0.51 dB less. A general rule
may be stated that for any given value of Ql/Qo, equal values
for Rag and Rd will result in the least possible IPL. In
this case, input and output return losses will be equal. It
appears to me that the parameters in lines 3 or 6 are probably
the ones to shoot for in most design calculations.
An effect I have observed is that one cannot simultaneously
attain a perfect impedance match at both input and output ports
in systems of the type shown in Fig. 1a (simultaneous conjugate
match). As shown on lines 4 and 5 in Table 2, Ragt and
Rdt can be arranged to provide a perfect match at one port,
but then the other port will be mismatched. Ro could be
replaced by a series resistor (not a real-world crystal radio set
anymore) and one would still not be able to arrange a simultaneous
impedance match. If circuit losses were represented by
proper values of both series and shunt resistances, it would
then be possible to attain a simultaneous perfect match at both
input and output.
An audio transformer is one passive device that has series
and shunt loss components. If it is so designed, it can
provide a simultaneous perfect impedance match at both input
and output when loaded by its designed-for load resistances
(ignoring reactance effects). If operated at any other
impedance level, such as doubling the source and load resistances,
simultaneous perfect input and output impedance matching cannot
be attained. This info is of mainly theoretical interest
for most crystal radio set applications except when one tries to operate
an audio transformer at considerably higher or lower source
and load resistance values than it was designed for. If
this is done, IPL is considerable increased compared to using
it the impedances for which it was designed.
A6. Audio transformer loss. See Articles
#1 and #5.
Part B: The Measurements.
Loss measurements at various frequencies on some components,
with the loss expressed as a parallel resistance (Rp) in parallel
with the capacitance of the component or the loss tangent, or equivalent
Q of the component:
-
B1: Q and equivalent parallel
loss resistance (Rp) of two variable capacitors vs frequency
when resonated with a 250 uH inductor, in parallel with a
total circuit stray circuit capacitance of 20 pF. Rp
and Q vs. frequency at four different capacitance settings
is also shown.
-
B2: Equivalent parallel loss
resistance of some DPDT slide switches.
-
B3: Loss tangent of some
coil form and sheet plastic (front panel) materials.
-
B4: Q of inductors and L/C resonators.
-
B5: High Q fixed value ceramic capacitors.
B1: Measurements at 520, 730,
943, 1300 and 1710 kHz, made on two different variable capacitors
used in crystal radio sets, are shown in the two graphs below.
Capacitor A is a 485 pF variable capacitor that was purchased
from Fair Radio Sales. It has a ceramic insulated
stator and silver plated brass plates with silver plated wiper
contacts. Capacitor B is a small 365 pF air-variable
capacitor purchased from the Xtal Set Society. Its plates
are made of aluminum and the stator support insulators are made
of a phenolic plastic. This capacitor is similar to those
sold by Antique Electronic Supply and others. See note at
the end of this section B1.
Fig. 2 shows Rp plotted against frequency, with
the capacitor adjusted at each frequency to a value 20 pf lower
than that required to resonate a 250 uH inductor. This
allows for a stray capacitance of 20 pF, in an actual circuit.
Any losses that may be in the stray capacitance are assumed assigned
to the inductor. The plot shows how, in actual practice,
the Rp of variable capacitors A and B vary when tuned across the
broadcast band. Fig. 3 shows how the Q of the total capacitance
(including the 20 pF stray capacitance) varies across the BC band
for each capacitor. Do not make the mistake,
when looking at the two graphs below, of thinking that they represent
Rp and Q of capacitors A or B vs. frequency, with the capacitor
set to a fixed capacitance. The capacitance is varied
as a function of frequency, along the horizontal axis of the graphs,
to a value that would resonate with a 250 uH inductor. Figs.
4 and 5 show Rp and Q of capacitor B as a function of frequency,
at four different fixed capacity settings (the frequency at each
capacitance setting is always a value that brings about resonance
with the 250 uH inductor).
|
|
Fig.4
|
Fig. 5
|
Discussion:
There are two main sources of loss in an air-dielectric
variable cap: 1) Loss in the dielectric of the stator insulators,
and 2) Resistive losses in the metal parts. Of course, there is
also the important very low-loss capacitor made from the air dielectric
between the plates. The losses in 2) include resistive loss
in the plates and in the wiper that connects the rotor to the
frame. Resistive loss in the plates is very small at low
frequencies, but increases with increasing frequency because of
skin effect.
Notice, that for capacitor B, Rp (vertical scale)
in Fig. 4 is about the same for all capacitance settings at the
lowest frequency plotted (310 kHz). Rp is approximately
constant over the full 14-365 pF capacitance range. This is
because we are varying a high Q air dielectric capacitor of very
low loss, in parallel with a low Q fixed capacitance made up of
the lossy phenolic stator supports. Most of Rp comes from
the loss in the phenolic. In Fig. 5, again at the low frequency
end of the horizontal axis, observe that Q is a direct linear
function of the overall capacitance, as it should be. The
Q of the air-variable part, taken alone, is much higher than that
of the capacitor made up of the phenolic insulators. The
main loss, here, comes from the approximately fixed shunt Rp provided
by the phenolic stator supports. At 14 pF the C from the
air cap is relatively low compared to that from the phenolic insulators.
At 365 pF the C from the air cap is much higher than that
from the phenolic supports.
Things change at higher frequencies. The
reactance of the capacitor drops. This, combined with
the series resistance of the plates and wiper now come into play
as an additional factor reducing the Q. If this series resistance
(Rs) were the only resistance affecting Q, the equation for Q
would be: Q=(reactance of the capacitor)/Rs. One can see
from this equation that the introduction of Rs makes the Q drop
when frequency increases. Up to now, the main loss
came from the parallel loss resistance of the phenolic supports.
The Q from a setting of 365 pF drops quite rapidly with
increase of frequency because of this series resistance. Skin
effect makes the effect worse by increasing Rs, the higher one
goes in frequency. Notice that at the low frequency end
of Fig. 5, Q is approximately proportional to the capacitance,
as it should be if the main loss is the fixed shunt resistance
Rp, coming from the phenolic stator insulators.
Fig. 5 shows an approximately constant Q (vs. frequency)
when the capacitor is set to 14 pF. The loss causing this
low, constant Q, comes from the loss tangent of the capacitor
formed from the phenolic stator insulators. At low frequencies,
Q increases when the capacitor is successively set to 114, 200
or 365 pF because engaging the plates adds a high Q air dielectric
capacitor component in parallel with the low Q capacitor formed
from the phenolic dielectric supports. As frequency increases,
with the capacitor set to 365 pF, one can see that the Q drops
at a faster rate than it does when set to 200 or 114 pF. This
is because Rs (being in series with the air capacitor, that dominates
at the 365 pF setting), acting with its lower capacitive reactance
results in a capacitor of lower Q (Q=reactance of air capacitor
at 365 pF/Rs).
In the Figs. 2 and 3, the capacitor is always set
for a circuit capacity value that resonates with 250 uH.
This means that at 520 kHz, the varicap is set so the circuit
capacity is 375 pF. At 1710 kHz, the circuit capacity is
set to 34.7 pF. Even though the capacitor Q goes from 19,000
to 9800 as frequency goes from 520 to 1710 kHz, Rp increases as
frequency increases because the circuit capacity must be reduced
from 375 to 34.7 pF to tune the tank from 520 to 1710 kHz (Rp=Q/(2*pi*f*C).
If the question is posed: 'Is it more important
to have ceramic insulated stators or silver plated plates
on a variable capacitor used in a BC band crystal radio set?', the answer
is that ceramic insulated stators are the way to go.
Capacitor A: The main practical conclusion
that can be taken from Fig .2 above is that Rp of capacitor A
is very high over the whole band and varies roughly proportionally
to frequency, over the frequency range of interest: 520-1710 kHz
In fact, Rp is so high that it will not contribute any
appreciable loss even when used in high performance crystal radio sets
using a high Q tank inductor. Another plus is that its Rp
increases with increasing frequency, further reducing any effect
on loaded Q, sensitivity or selectivity at the high end of the
band. The silver plating on the brass plates is beneficial
because the resistivity of silver is about 25% of that of brass.
Practically speaking, the silver should have little effect
on the operation of a crystal radio set in the BC band, but short wave
is another matter. The Q of the capacitor drops at higher
frequencies, especially when set to a high capacitance value.
Silver plating can materially improve performance at higher
frequencies by providing a needed higher Q. Be aware that
some capacitors are made with cadmium plated brass plates. Silvery-whitish
colored cadmium has a resistivity 4.6 times that of silver. This
higher resistivity will somewhat reduce Q at high frequencies
and at high capacity settings. Some people mistakenly assume,
because of the silvery-whitish color, that cadmium plated plates
are really silver plated.
Capacitor B: One can see from Fig.
2 that Rp of capacitor B varies approximately inversely with frequency.
The DLF (dielectric loss factor) of the phenolic stator support
insulators is the main cause of this loss, over the whole frequency
band. Towards the high end of the band, some loss is contributed
by the series resistance in the capacitor plates, the rotor shaft
wiper contact and skin effect. This loss effect is greater
than that in capacitor A because the resistivity of the aluminum
plates is 1.7 times that of silver. Practically speaking,
this effect is minimized at the high end of the band because the
plates are mostly disengaged. The Rp of the capacitor will
have its greatest effect in reducing sensitivity and selectivity
at the high frequency end of the band because that is where its
value is the the lowest. See note at the end of this Article.
The usual crystal radio set uses shunt capacitor tuning
with a fixed tank inductance. This configuration causes
the tank reactance to be highest at the high end of the band,
thus further reducing loaded Q, sensitivity and selectivity, for
a given value of Rp. At the low and medium frequency parts
of the band, Rp of the capacitor is so high that its effect is
small in many crystal radio sets. Highest performance crystal
radio sets made with high Q inductors, with careful attention to
impedance matching may experience a noticeable reduction in sensitivity
and selectivity at the high end of the band when using
this or other capacitors using phenolic stator insulation.
This is because the Rp of the capacitor becomes comparable to
the higher equivalent Rp of the high Q tank inductor at the high
end of the band as compared to its value at the low end.
To clarify this, an ideal condition would exist if Rp of the capacitor
and inductor were infinite. If this were to be the case,
and good impedance matching of antenna-ground system to tank,
diode to tank and headphones to diode existed, all of the power
intercepted by the antenna-ground system would be delivered to
the headphones and maximum sensitivity would occur. Any
loss present in capacitors, inductors or transformers reduces
sensitivity. In this discussion we are dealing primarily
with losses in the resonating capacitor and tank inductor, both
referred to as their respective values of Rp. The higher
the value of these Rps, compared to the transformed antenna-ground
source resistance across which they appear, the lesser the loss
they cause. One approach to counter the effect the drop
in Rp as frequency increases is to change to two-step inductive
tuning by dividing the band into two sections as described in
Article #22.
Technical Note: At any frequency, a real world capacitor
of value C1 having a Q of Q1 (Q1>10), can be quite accurately
modeled as a series combination of an ideal no-loss capacitor
of value C1 and a resistor (Rs) equal to: (reactance of C1)/Q1.
Alternatively, at any frequency, a capacitor of value C1, having
a Q of Q1 (Q1>10) can be modeled as a parallel combination of
an ideal capacitor of value C1 and a resistor (Rp). The
resistor, Rp in this case, has a value of:: (reactance of C1)*Q1. Note that since
capacitor reactance is a function of frequency, the value of the
resistor will, in general, vary with frequency. In crystal
radio set design it is sometimes convenient to model the tuning capacitor
loss as a parallel resistor, other times as a series resistor.
Credit must go to Bill Hebbert for making the time
consuming, difficult, precision measurements required for Figs.
2-5.
B2: Slide switches used in the
Crystal Radio Set described in Article #22 have dielectric losses, as
do all switches. To get a handle on this loss, samples of
several different types of DPDT switch were measured at 1500 kHz.
Each switch except the last three, below, were wired as a SPDT
unit by paralleling the two sections. The Q of the capacitance
appearing across the open contacts was then measured and Rp was
calculated. Rp usually varies approximately inversely with
frequency and therefore causes more loss at the high end of the
band than at the low end. Contact resistance of all switches
was found to be very low. It is unknown how well that characteristic
will hold up over time. Note the extremely low loss of the
Switchcraft 56206L1. The loss is so low that this switch
is overkill in most crystal radio set applications. For applications
in which the crystal radio set builder wants to use the lowest loss
DPDT slide-switch available, this switch is the best I have found.
Table 3 - Equivalent parallel Loss
Resistance (Rp) caused by the
loss tangent of the dielectric of some
DPDT Slide Switches at 1.5 MHz. |
Brand name, model number
and Size of switch |
Rp in
Megohms |
ARK-LES (std. size) |
2.3
|
Radio Shack 275-403A (std. size) |
3.5
|
Radio Shack 275-407A (sub mini) |
4.1
|
Stackpole S7022X (std. size) |
4.9
|
Switchcraft 46206EE-6 (std. size) |
5.1
|
Radio Shack 275-327B (mini) |
5.7
|
CW (mini) |
6.0
|
Switchcraft 46206LR (std. size) |
6.5
|
C&K L202-1 (mini) |
6.8
|
Switchcraft 56206L1 (mini) |
13.3
|
Radio Shack 275-327B connected
as SW3 in Article #22 (HI band) |
4.1
|
Switchcraft 56206L1 connected as
SW6 in Article #22 |
9.4
|
Radio Shack 275-327B connected as
SW6 in Article #22, See Graph--->> |
3.5
|
|
|
Rotary selector switches using ceramic insulation should have very low loss, even lower than the Switchcraft 56206L1.
Quality switches using brown phenolic insulation probably have
losses similar to slide switches using similar material. I would expect that the slope of the Rp vs. frequency graph of
the other slide switches to be the similar that shown above.
B3: The loss tangent
of an insulating material is the reciprocal of the Q of a capacitor
made of that material. Some of the insulating materials
listed below are used as front panels, detector stand bases, wire
insulation and coil forms in crystal radio sets. A capacitor formed
by the use of one of these materials, connected across a high
impedance point and ground, will contribute a loss proportional
to the loss tangent and capacitance.
Table 4 - Loss Tangent of some
Insulating Materials
used in Crystal Radio Sets, Measured at
1.5 MHz.
Dielectric Material
|
Loss tangent
|
Q of a capacitor using the material
|
PVC as used in coil form |
0.017
|
59
|
PVC wire insulation |
0.03
|
33
|
High-impact styrene coupler from
Genova Mfg. Co. (opaque white) |
0.0017
|
590
|
Polypropylene 1.5" diameter drain pipe from Genova Mfg. Co. |
0.0022 |
452 |
High impact styrene sheet, 0.1" thick
(opaque white and flexible) |
0.0023
|
430
|
Plexiglas 0.115" thick |
0.016
|
55
|
FR-4 PCB material 1/16' thick |
0.027
|
37
|
Black 3/16" Condensite panel,
brand name "Celoron"
(Bakelite, new
old-stock radio panel from the '20's) |
0.035 (@ 0.8 MHz)
|
29
|
ABS styrene (black) |
0.010 |
100 |
Garolite (black) |
0.033 |
30 |
ABS styrene (light beige) |
0.020 |
50 |
HDPE (milky white) |
0.0009 |
1120 |
Polystyrene (light brown opaque) |
0.0032 |
320 |
Note: GE's version of polycarbonate is called Lexan. A review GE's spec. sheets of various grades of Lexan show loss tangent values at 1.0 MHz ranging between 0.006 and 0.026 . Many grades are specified 0.01.
B4: Please see Parts 10 and 11 of Article #26 as well as Table 4. Also see Table 2 of Article #22 and Article #29.
B5: Sometimes, when working with high Q tank circuits, a need pops up for a fixed capacitor with a value between say, 100 and 1000 pF that will not degrade overall circuit Q. Generic NPO disc capacitors in that range usually have a Q of around 2000-3000 at 1 MHz. Table 4 shows some caps having higher Q values. The only downside to the high Q caps is that they are SMD types and require some skill when soldering pigtail leads to them to easy connecting to one's circuit. The capacitors were measured singly, in parallel or in series, aiming for values approximating 500 pF. This was for convenience in measurement. I used solid tinned copper wire having a diameter of about 0.010" for my pig-tails. The source for the strands was a piece of stranded hook-up wire.
Table 4 - Q of easily available capacitors in the 100 - 1000 pF range |
|
Type |
Value
in pF |
Voltage |
Q at about 920 kHz |
Mfg |
Mfg. part number |
1 |
Polypropylene |
Two 1k
in series=500 |
630 |
2,000 |
Xicon (Mouser) |
1431-6102K |
2 |
Polystyrene |
One 470 |
50 |
6,200 |
Xicon (Mouser) |
23PS147 |
3 |
Generic NPO disc |
One 220 |
500 |
2,800 |
3/8" dia. |
NA |
4 |
Multilayer, hi Q SMD |
Two 1k in series=500 |
50 |
8,000 |
Murata |
ERB32Q5C1H102JDX1L |
5 |
Multilayer, hi Q SMD |
One 470 |
100 |
18,000 |
Murata |
ERB32Q5C2A471JDX1L |
6 |
Multilayer, hi Q SMD |
Two 220 in parallel=440 |
200 |
20,000 |
Murata |
ERB32Q5C2D221JDX1L |
7 |
Multilayer, hi Q SMD |
Two100 in parallel=200 |
500 |
40,000 |
Murata |
ERB32Q5C2H101JDX1L |
Note: Solder flux contamination on a dielectric
is the enemy of high Q because it usually provides a resistive
leakage path. If one gets solder flux on the insulation
of a variable cap or switch, remove it with a commercial flux
remover. This is important when a DX crystal radio set is involved.
|